Beaulieu series approach to optimal UMTS RACH preamble detection estimation

ABSTRACT

A method for preamble detection in mobile unit to base unit wireless telephony sets a preamble detection threshold based upon a Beaulieu series computation dependent upon preamble correlation data. This preamble detection threshold adjusts for noise by assuming the noise is additive White Gaussian noise (AWGN) with a known variance. The method determines the threshold for achieving a probability of false detection of the preamble from noise input of less than 0.001.

CLAIM OF PRIORITY

This application claims priority under 35 U.S.C. 119(c) from U.S. Provisional Application 60/562,867 filed Apr. 16, 2004.

TECHNICAL FIELD OF THE INVENTION

The technical field of this invention is determining the detection threshold for preamble detection for mobile telephone users for connection to a UMTS base-station receiver.

BACKGROUND OF THE INVENTION

One major challenge in RACH preamble detection is the determination of the detection threshold. A power estimator (PE) can be used to estimate the input noise variance. This PE based technique assumes that the signal-to-noise ratio is low enough so that what is measured by PE is just the interference plus the noise level. This approach works well and converges in a few slots in low-to-moderate signal-to-noise ratio (SNR) environments. Dynamic range adjusters (sliders) in PE may accidentally not be set the same as those in preamble detector (PD). The software complexity increases if we need to pass PE values for use in PD and path monitor (PM) algorithms. Finally, in high SNR scenarios, the PE noise estimate will be strongly influenced by the preamble itself.

Another existing approach to this problem configures a preamble correlation hardware block with a scrambling code that is not used by the mobiles in the particular cell or sector of concern. This requires additional hardware that is used wastefully. Furthermore, the estimation of the detection threshold is still not resolved.

SUMMARY OF THE INVENTION

This invention uses statistics derived with Beaulieu's method to determine the detection threshold assuming input additive white Gaussian noise (AWGN) with known variance. Prior methods use the PE to estimate the input noise variance. However, it is preferable to use the PD itself to characterize the noise variance. For one, this would capture the impact of the PD hardware such as bit-widths. In high SNR scenarios, the orthogonality of the PRACH signatures may allow a more accurate estimate of the noise.

In a typical wireless 3G scenario, there will be free signatures. Thus for a given PRACH scrambling code not all of the 16 signatures will be assigned to users. This invention is an approach to estimate the input noise variance directly from PD outputs if signatures are free. An alternative technique based on PE may be used when no signatures are free. Simulations show that the error using this method is small even using just a few PD output observations. There is a 95% confidence that the fractional error is 2% using just 6 sets of PD outputs.

The PE based approach of the prior art assumes that the signal-to-noise ratio is low enough so that what is measured by PE is just the interference plus noise. This inherent assumption limits the operation range of the preamble detector. In this invention the PE and the preamble correlations are computed separately. Therefore results from PE need to be mapped to whatever configuration is used for the preamble correlations. Convergence speed of the threshold statistic computed according to this invention is much faster than the PE based solution.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other aspects of this invention are illustrated in the drawings, in which:

FIG. 1 illustrates a block diagram of a preamble detector;

FIG. 2 illustrates a graph of the probability density function versus z for W=512 and N_(nca)=2 and several values of i;

FIG. 3 illustrates a graph of the fractional error percentage versus access slots to produce a 95% that the factional error is less than the ordinate;

FIG. 4 illustrates a graph of the probability of false detection of the preamble P_(FA) for several values of L={circumflex over (N)}_(nca) assuming ${\hat{W} = {{512\quad{and}\quad\frac{\sigma_{n,{nca}}^{2}}{2}} = 1}};$

FIG. 5 illustrates a graph of the probability that the estimate has fractional error of 1% versus the number of samples N;

FIG. 6 illustrates a graph of the probability that the estimate has fractional error of 5% versus the number of packets N;

FIG. 7 illustrates a graph of the base 10 logarithm of the number of samples N versus α needed to achieve 95% confidence;

FIG. 8 illustrates a graph of the probability that the fractional error is less than α versus α for 4096 samples; and

FIG. 9 illustrates the steps of this invention in flow chart form.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

Preamble detection is the first step in the wireless telephone base station sorting out the signals from plural wireless telephone users. The base station must determine which received radio transmission corresponds to which user. This determination is necessary to apply the proper signal conditioning such as echo cancellation. This determination also permits the base station to route a recovered voice stream to the counterpart party to a particular wireless telephone user.

Setting a preamble detection threshold is crucial proper detection. A too high threshold would prevent detection of some valid preambles corresponding to valid radio transmissions. If the threshold is set too low, then there is an increased chance of a false positive preamble detection from received noise. Setting the preamble detection threshold thus involves a compromise between missing too many proper transmissions and making too many false determinations.

FIG. 1 illustrates an example preamble detector (PD) that supports the detection of preambles on the PRACH. Within a predefined search window, the PD stores the 16 largest detection results and the associated offsets for each of the 16 signatures. An offset step size of ½ or 1 chip can be used. The PD passes these results to a chip rate assist (CRA) digital signal processor (DSP). The chip rate assist DSP determines if a preamble is present, acknowledges its detection and programs the finger despreader (FD) and path monitor (PM) accordingly.

Input buffer 101 receives the detected radio frequency signals as 8-bit I/Q signals. Input buffer 101 preferably has the capacity to store 1408 samples of each of 12 data streams of 2 times OSF 8-bit I/Q data per sample. Input buffer 101 supplies 64 chips to correlator 102. Correlator 102 correlates the 64 chips from input buffer 101 with PRACH scrambling codes from code generator 103. Correlator 102 preferably performs 2048 simultaneous 64-chip correlations.

Suppose x_(n) and y_(n) model the respective in-phase and quadrature input of additive White Gaussian noise (AWGN) samples, each having zero mean with variance $\frac{\sigma_{n}^{2}}{2}.$ Then let X_(n,ca) and Y_(n,ca) model the coherent accumulation of x_(n) and y_(n) respectively over N_(ca) chips. X_(n,ca) and Y_(n,ca) are also Gaussian with mean zero and variance $\frac{\sigma_{n,{co}}^{2}}{2} = {N_{ca}{\frac{\sigma_{n}^{2}}{2}.}}$

The PD forms the absolute value of the coherent accumulation via rotator 104, coherent accumulator 105, coherent scratch memory 106 and Hadamard transformer 107. Rotator 104 preferably has a range of ±30 kHz in steps of 60 Hz. Coherent accumulator 105 employs coherent scratch buffer 106 to store intermediate results. Coherent scratch buffer 106 preferably has a capacity of 32,76832-bit data words. Coherent accumulator 105 provides an output dynamic range selection to Hadamard transform 107. The result Z_(n,ca)={square root}{square root over (X_(n,ca) ²+Y_(n,ca) ²)} has Rayleigh distribution with mean $m_{z_{n,{ca}}} = {{\frac{\sigma_{n,{ca}}^{2}}{2}\sqrt{\frac{\pi}{2}}} = {\frac{N_{ca}\sigma_{n}^{2}}{2}\sqrt{\frac{\pi}{2}}}}$ and variance $\sigma_{z_{n,{ca}}}^{2} = {{\left( {2 - \frac{\pi}{2}} \right)\frac{\sigma_{n,{ca}}^{2}}{2}} = {\left( {2 - \frac{\pi}{2}} \right){\frac{N_{ca}\sigma_{n}^{2}}{2}.}}}$

The PD then non-coherently combines N_(n,ca) coherent packets Z_(n,ca) via non-coherent accumulator 109 and non-coherent scratch buffer 110. As coherent scratch buffer 106, non-coherent scratch buffer 110 preferably has a capacity of 32,768 32-bit words. This forms $Z_{n,{nca}} = {{\sum\limits^{N_{nca}}Z_{n,{ca}}} = {\sum\limits^{N_{nca}}{\sqrt{X_{n,{ca}}^{2} + Y_{n,{ca}}^{2}}.}}}$ It is convenient to normalize the non-coherent output by $\frac{\sigma_{n,{ca}}}{\sqrt{2}}$ yielding ${\overset{\sim}{Z}}_{n,{nca}} = {\frac{\sqrt{2}\quad Z_{n,{nca}}}{\sigma_{n,{ca}}}.}$ The results of non-coherent accumulator 109 are sorted via sorter 111 and output via output buffer 112. Output buffer 112 preferably has a capacity of 2048 32-bit words.

The next section describes a Beaulieu series approach to determine the statistics of {tilde over (Z)}_(n,nca). Let X_(i), i=1 to L be independent Rayleigh variables with probability density function (pdf) given by: ${f_{x_{i}}\left( x_{i} \right)} = \left\{ \begin{matrix} {{\frac{x_{i}}{\sigma_{i}^{2}}{\mathbb{e}}^{{{- x_{i}^{2}}/2}\sigma_{i}^{2}}},} & {x \geq 0} \\ {0,} & {otherwise} \end{matrix} \right.$ where (2−π/2)σ_(i) ² is the variance of the distribution.

Denote the sum of the L Rayleigh variables by $X = {\sum\limits_{i = 1}^{L}X_{i}}$ the complementary distribution G_(x)(x), cumulative distribution F_(x)(x), and probability density function f_(x)(x). Then: $\begin{matrix} {{{G_{x}\left( {ɛ\quad L} \right)} = {\frac{1}{2} + {\frac{2}{\pi}{\sum\limits_{\underset{n\quad{odd}}{n = 1}}^{\infty}\frac{\left( A_{in} \right)^{L}{\sin\left( {L\quad\theta_{in}} \right)}}{n}}}}}{with}} & (1) \\ {A_{in} = \sqrt{\left\lbrack {{{}_{}^{}{}_{}^{}}\left( {1,\frac{1}{2},\frac{{- n^{2}}\omega^{2}\sigma_{i}^{2}}{2}} \right)} \right\rbrack^{2} + {\frac{\pi}{2}n^{2}\omega^{2}\sigma_{i}^{2}{\mathbb{e}}^{{- n^{2}}\omega^{2}\sigma_{i}^{2}}}}} & (2) \\ {\theta_{in} = {\tan^{- 1}\left\{ \frac{\begin{matrix} {{\frac{\pi}{2}n^{2}\omega^{2}\sigma_{i}^{2}{\mathbb{e}}^{{- n^{2}}\omega^{2}\sigma_{i}^{2}}{\cos\left( {n\quad{\omega ɛ}} \right)}} -} \\ {{{{}_{}^{}{}_{}^{}}\left( {1,\frac{1}{2},\frac{{- n^{2}}\omega^{2}\sigma_{i}^{2}}{2}} \right)}{\sin\left( {n\quad{\omega ɛ}} \right)}} \end{matrix}}{\begin{matrix} {{{{{}_{}^{}{}_{}^{}}\left( {1,\frac{1}{2},\frac{{- n^{2}}\omega^{2}\sigma_{i}^{2}}{2}} \right)}{\cos\left( {n\quad{\omega ɛ}} \right)}} +} \\ {\frac{\pi}{2}n^{2}\omega^{2}\sigma_{i}^{2}{\mathbb{e}}^{{- n^{2}}\omega^{2}\sigma_{i}^{2}}{\sin\left( {n\quad{\omega ɛ}} \right)}} \end{matrix}} \right\}}} & (3) \end{matrix}$ where ₁F₁(.,.,.) is the confluent hypergeometric function and X is a term affecting the convergence rate of G_(x)(εL).

Denote the i^(th) largest normalized PD output in a single antenna search window of W as ({tilde over (Z)}_(n,nca))_(i). Ding, C-S, et al., “Statistical Estimation of the Cumulative Distribution Function for Power Dissipation in VLSI Circuits,” Proceedings of the 34^(th) Design Automation Conferences, Jun. 9-13, 1997, pp. 371-376 then shows that these ordered statistics have a power dissipation function given by: $\begin{matrix} {{p\left( {\left( {\overset{\sim}{Z}}_{n,{nca}} \right)_{i} = z} \right)} = {\frac{n!}{{\left( {i - 1} \right)!}{\left( {n - 1} \right)!}}\left( {F_{{\overset{\sim}{Z}}_{n,{nca}}}(z)} \right)^{n - i}\left( {G_{{\overset{\sim}{Z}}_{n,{nca}}}(z)} \right)^{i - 1}{f_{{\overset{\sim}{Z}}_{n,{nca}}}(z)}}} & (4) \end{matrix}$

p(({tilde over (Z)}_(n,nca))_(i)=z) and the expected value E(({tilde over (Z)}_(n,nca))_(i))=ξ_(i,W,N) _(nca) can be derived numerically for any value of W, N_(nca) and i by setting σ_(i) ²=1 in Equations 1 to 3 and using the results in Equation 4. FIG. 2 illustrates the probability density function p(({tilde over (Z)}_(n,nca))_(i)=z) versus z for W=512 and N_(nca)=2 and several values of i. Table 1 provides ξ_(i,W,N) _(nca) for W=512 and W=1024 and several values of N_(nca). TABLE 1 W = 512 W = 1024 i N_(nca) = 1 N_(nca) = 2 N_(nca) = 4 N_(nca) = 8 N_(nca) = 1 N_(nca) = 2 N_(nca) = 4 N_(nca) = 8 1 3.667330 5.778324 9.477385 16.161117 3.864348 6.038004 9.821462 16.620744 2 3.376443 5.394726 8.968832 15.481237 3.585999 5.671316 9.335858 15.972554 3 3.221183 5.189702 8.696589 15.116448 3.438471 5.476691 9.077762 15.627374 4 3.113599 5.047540 9.507629 14.862889 3.336718 5.342364 8.899480 15.388651 5 3.030589 4.937802 8.361650 14.666792 3.258494 5.239050 8.762274 15.204764 6 2.962647 4.847953 8.242052 14.505987 3.194668 5.154723 8.650226 15.054484 7 2.904924 4.771599 8.140357 14.369148 3.140590 5.083256 8.555224 14.926984 8 2.854606 4.705025 8.051642 14.249691 3.093565 5.021097 8.472561 14.815983 9 2.809909 4.645877 7.972784 14.143442 3.051888 4.965997 8.399259 14.717503 10 2.769633 4.592568 7.901681 14.047586 3.014412 4.916442 8.333313 14.628864 11 2.732926 4.543978 7.836843 13.960130 2.980325 4.871362 8.273304 14.548171 12 2.699166 4.499281 7.777177 13.879610 2.949035 4.829974 8.218192 14.474035 13 2.667882 4.457857 7.721858 13.804921 2.920090 4.791685 8.167192 14.405404 14 2.638708 4.419221 7.670244 13.735204 2.893142 4.756035 8.119696 14.341466 15 2.611355 4.382992 7.621829 13.669780 2.867913 4.722662 8.075221 14.281576 16 2.585590 4.348863 7.576204 13.608102 2.844191 4.691273 8.033382 14.225217 The unnormalized preamble output (Z_(n,nca)) has a mean ${E\left( \left( Z_{n,{nca}} \right)_{i} \right)} = {\frac{\sigma_{n,{nca}}}{\sqrt{2}}{\xi_{i,W,N_{nca}}.}}$ With N_(B) preamble detection attempts we can thus form an estimate of σ_(n,ca) using: $\begin{matrix} {{\hat{\sigma}}_{n,{ca}} = {\frac{\sqrt{2}}{16N_{B}}{\sum\limits_{j = 1}^{N_{B}}{\sum\limits_{i = 1}^{16}\frac{\left( Z_{n,{nca}} \right)_{j,i}}{\xi_{i,W,N_{nca}}}}}}} & (5) \end{matrix}$

Preamble detector simulations show that such an estimator converges very rapidly without bias. FIG. 3 illustrates a graph of the fractional error percentage versus access slots to produce a 95% that the factional error is less than the ordinate. For example, assuming L=2 and W=512, FIG. 3 shows that N_(B)=6 access slots are enough for 95% confidence that the standard deviation estimate is within 2% of the actual. N_(B)=35 yields a 95% confidence that the estimate is within 1%.

The performance requirement of RACH for preamble detection is determined by the two parameters probability of false detection of the preamble P_(FA) and the probability of detection of preamble P_(D). The performance is measured by the required energy-per-chip to noise power spectral density ratio, E_(c)/N₀ at a probability of detection, P_(D), of 0.99 and 0.999. P_(FA) is defined as a conditional probability of erroneous detection of the preamble when input is only noise and interference. P_(D) is defined as conditional probability of detection of the preamble when the signal is present. P_(FA) should be 10⁻³ or less. Only one signature is used and it is known by the receiver.

P_(FA) depends on the diversity combining method. Consider “selection diversity” and “diversity combining.” With selection diversity, the length 16 sorted lists are taken from each antenna and concatenated. If any of the resulting 32 values exceeds the threshold, a detection is declared. With selection diversity, the equivalent search window is twice the search window used in each antenna. With diversity combining, the sorted length 16 lists are taken to see if particular offsets are in both lists. If so, the search values are added. A new list of length between 16 and 32 is formed with the combined results. If a particular offset is only observed in a single antenna, it is added as is to the new list. Each of the resultant samples is based on the single antenna search window. However, each sample, in the worst case assuming only noise is present, represents the non-coherent accumulation over twice the number of coherent packets used to generate the single antenna results. Table 2 generalizes the adjustments needed with multiple antennas. TABLE 2 Equivalent Equivalent Number of Diversity Diversity Search Coherent Method Method Window W Packets N_(nca) Details Diversity W M N_(nca) Combine Combining results with common offsets Selection MW N_(nca) Concatenate Diversity lists W = single antenna search window size in offsets N_(nca) = single antenna number of coherent packets

We can use Beaulieu's approach to determine P(Z_(n,nca)) for a given value of {circumflex over (N)}_(nca) and then P_(FA) from: P_(FA) = P(all  Z_(n, nca) ≤ τ) = P(Z_(n, nca) ≤ τ)^(Ŵ) FIG. 4 illustrates a graph of the probability of false detection of the preamble P_(FA) versus Z_(n,nca) for several values of L={circumflex over (N)}_(nca) assuming $\hat{W} = {{512\quad{and}\quad\frac{\sigma_{n,{nca}}^{2}}{\sqrt{2}}} = 1.}$ FIG. 4 suggests that for P_(FA)<0.001, we'd need a detection threshold of 7.95, 12.40, and 20.09 for L={circumflex over (N)}_(nca)=2, 4 and 8, respectively. Since Z_(n,nca) scales directly with σ_(n,ca), the results of FIG. 4 can be used for any value of σ_(n,ca) assuming a constant value of Ŵ. The detection threshold is simply ${7.95\frac{\sigma_{n,{ca}}}{\sqrt{2}}},{{12.40\frac{\sigma_{n,{ca}}}{\sqrt{2}}\quad{and}\quad 20.09\frac{\sigma_{n,{ca}}}{\sqrt{2}}\quad{for}\quad L} = 2},4,{{and}\quad 8},$ 4, and 8, respectively, for instance in the FIG. 4 scenario.

In general, the detection threshold τ, is given by: $\begin{matrix} {\tau = {C_{{\hat{N}}_{nca},\hat{W}}\frac{\sigma_{n,{ca}}}{\sqrt{2}}}} & (6) \end{matrix}$

where C_({circumflex over (N)}) _(nca) _(,Ŵ) is a fixed constant depending on {circumflex over (N)}_(nca) and Ŵ, for example given in Table 3 for Ŵ=512 and Ŵ=1024 and {circumflex over (N)}_(nca) is 2, 4, 6 and 8. TABLE 3 W = 512 W = 1024 N_(nca) = 1 N_(nca) = 2 N_(nca) = 4 N_(nca) = 8 N_(nca) = 1 N_(nca) = 2 N_(nca) = 4 N_(nca) = 8 5.104674 7.769179 12.155090 9.756381 5.221810 7.952168 12.398603 20.086897 Substituting Equation 5 into Equation 6 gives an estimate of the threshold: $\begin{matrix} {\hat{\tau} = {\frac{1}{16N_{B}}{\sum\limits_{j = 1}^{N_{B}}{\sum\limits_{i = 1}^{16}\frac{{C_{{\hat{N}}_{nca},\hat{W}}\left( Z_{n,{nca}} \right)}_{j,i}}{\xi_{i,W,N_{nca}}}}}}} & (7) \end{matrix}$ The accuracy of this estimate is as accurate as the estimate σ_(n,ca). For PD outputs from 10 access slots, we should have a great deal of confidence that the fractional error of our estimate is very small.

Roughly 10 samples (access slot, signature pairs) are required for statistical confidence. Thus, the following approach is proposed. For initialization of the first 10 samples t, form the threshold estimate using all t samples: $\hat{\tau} = {\frac{1}{16t}{\sum\limits_{j = 1}^{t}{\sum\limits_{i = 1}^{16}\frac{{C_{{\hat{N}}_{nca},\hat{W}}\left( Z_{n,{nca}} \right)}_{j,i}}{\xi_{i,W,N_{nca}}}}}}$ Upon reaching equilibrium, form the instantaneous threshold estimate: $\hat{\tau} = {\frac{1}{16}{\sum\limits_{i = 1}^{16}\frac{{C_{{\hat{N}}_{nca},\hat{W}}\left( Z_{n,{nca}} \right)}_{j,i}}{\xi_{i,W,N_{nca}}}}}$ and insert into infinite impulse response (IIR) filter: {circumflex over (τ)}_(k)=(1−α){circumflex over (τ)}_(k-1)+α{circumflex over (τ)} where α=¼.

In an alternative to this invention, the power estimate could be used to estimate the standard deviation of the input interference in the preamble detector threshold selection. Let x_(n) and y_(n) be normal variables with zero mean and variance σ². It is well known that the sample mean ${Z = \frac{{\sum\limits_{n = 1}^{N}x_{n}^{2}} + y_{n}^{2}}{N}},$ is the ML estimate (unbiased) of 2σ².

The term x_(n) ²+y_(n) ² is the chi-squared with mean 2σ² and variance 4σ⁴. Assuming that N is large, according to the central limit theorem Z is approximately normal with mean 2σ² and variance (4/N)σ⁴. The probability that Z is within a certain fractional error from the mean can then be estimated by: $\begin{matrix} {{p\left( {{{Z - {2\sigma^{2}}}} < {\alpha 2\sigma}^{2}} \right)} = {1 - {2{p\left( {{Z - {2\sigma^{2}}} > {\alpha 2\sigma}^{2}} \right)}}}} \\ {= {1 - {2{p\left( {Z > {\left( {\alpha + 1} \right)2\sigma^{2}}} \right)}}}} \\ {= {1 - {2{Q\left( \frac{{\left( {\alpha + 1} \right)2\sigma^{2}} - {2\sigma^{2}}}{\sqrt{\frac{4}{N}\sigma^{4}}} \right)}}}} \\ {= {1 - {2Q\text{(}\sqrt{N}\alpha\text{)}}}} \end{matrix}$ For instance, FIGS. 5 and 6 show the probability that the estimate has fractional error of 1 and 5%, respectively (e.g., for α=0.01 and α=0.05).

FIG. 7 illustrates the value of N on a logarithmic scale required to achieve 95% confidence, i.e., p(|Z−2σ²|<α2σ²)=0.95, as a function of α.

FIG. 8 illustrates the probability that the fractional error is less than a as a function of α for N=4096.

Applying the above results to estimate the input interference level, we set a goal of a confidence of 95% that the noise variance estimate has a fractional error relative to the actual within 1 to 2%. Simulations show that for N=4096, the error is within 1.2% only 56% of the time. An N of 26677 is required for 95% confidence. An N of 4096 yields 95% confidence that error is within 3%, 99% within 4%, and 99.9% within 5%. Given N=9603, 95% of the time the error is within 2%. For N=38414, 95% of the time the error is within 1%. Averaging over about 20000 samples or about 8 slots on generation of the noise estimate yields 95% confidence that the error is within 1.4%. Employing a number of samples of this order is advisable.

Implementation with power estimation is similar to that described above. Initialization of the first 80 256-chip PE outputs N, includes forming sample mean from PE outputs: $\hat{Z} = \frac{{\sum\limits_{n = 1}^{N}x_{n}^{2}} + y_{n}^{2}}{N}$ and forming the detection threshold estimate from: $\hat{\tau} = {C_{{\hat{N}}_{nca},\hat{W}}\sqrt{\frac{4096}{N_{nca}}\hat{Z}}}$ Upon equilibrium take the PE output {circumflex over (Z)} and feed into infinite impulse filter (IIR) filter: {tilde over (Z)} _(k)=(1−α){tilde over (Z)} _(k-1) +α{tilde over (Z)} where α= 1/16. The detection threshold is formed from: $\hat{\tau} = {C_{{\hat{N}}_{nca}\hat{,W}}\sqrt{\frac{4096}{N_{nca}}\overset{\sim}{Z}}}$ 

1. A method of radio frequency message preamble detection comprising the steps of: receiving in-phase and quadrature input samples of the radio frequency message; coherently accumulating a first predetermined number of said in-phase and quadrature input samples; forming an absolute value of the coherently accumulated input samples; non-coherently combining a second predetermined number of said absolute values of said coherently accumulated input samples; selecting a preamble detection threshold τ according to the equation: $\hat{\tau} = {\frac{1}{16N_{B}}{\sum\limits_{j = 1}^{N_{B}}{\sum\limits_{i = 1}^{16}\frac{{C_{{\hat{N}}_{nca},\hat{W}}\left( Z_{n,{nca}} \right)}_{j,i}}{\xi_{i,W,N_{nca}}}}}}$ where: N_(B) is a number of preamble detection attempts; C_({circumflex over (N)}) _(nca) _(,Ŵ) is a fixed constant depending on {circumflex over (N)}_(nca) and Ŵ; {circumflex over (N)}_(nca) is the equivalent number of coherent packets; Ŵ is the equivalent search window; (Z_(n,nca))_(j,i) is the non-coherently combined values; and ξ_(i,W,N) _(nca) is the expected value of (Z_(n,nca))_(j,i); and detecting a preamble in the radio frequency message employing said selected preamble detection threshold τ.
 2. The method of claim 1, wherein: said constant C_({circumflex over (N)}) _(nca) _(,Ŵ) is selected from the table: W = 512 W = 1024 N_(nca) = 1 N_(nca) = 2 N_(nca) = 4 N_(nca) = 8 N_(nca) = 1 N_(nca) = 2 N_(nca) = 4 N_(nca) = 8 5.104674 7.769179 12.155090 9.756381 5.221810 7.952168 12.398603 20.086897

according to the selected values of W and N_(nca).
 3. The method of claim 1, wherein: said step of selecting said preamble detection threshold τ further includes selecting said preamble detection threshold τ for a first 10 samples according to the equation: $\tau = {\frac{1}{16t}{\sum\limits_{j = 1}^{t}{\sum\limits_{i = 1}^{16}\frac{{C_{{\hat{N}}_{nca},\hat{W}}\left( Z_{n,{nca}} \right)}_{j,i}}{\xi_{i,W,N_{nca}}}}}}$ forming an instantaneous preamble detection threshold {circumflex over (τ)} for following samples according to the equation: $\hat{\tau} = {\frac{1}{16}{\sum\limits_{i = 1}^{16}\frac{{C_{{\hat{N}}_{nca},\hat{W}}\left( Z_{n,{nca}} \right)}_{j,i}}{\xi_{i,W,N_{nca}}}}}$ modifying a preamble detection threshold {circumflex over (τ)}_(k) according to the following infinite impulse response (IIR) filter: {circumflex over (τ)}_(k)=(1−α){circumflex over (τ)}_(k-1)+α{circumflex over (τ)} where α=¼.
 4. A method of radio frequency message preamble detection comprising the steps of: receiving in-phase and quadrature input samples of the radio frequency message; coherently accumulating a first predetermined number of said in-phase and quadrature input samples; forming an absolute value of the coherently accumulated input samples; non-coherently combining a second predetermined number of said absolute values of said coherently accumulated input samples; selecting a preamble detection threshold {circumflex over (τ)} according to the equation: $\hat{\tau} = {C_{{\hat{N}}_{nca}\hat{,W}}\sqrt{\frac{4096}{N_{nca}}\overset{\sim}{Z}}}$ where: C_({circumflex over (N)}) _(nca) _(,Ŵ) is a fixed constant depending on {circumflex over (N)}_(nca) and Ŵ; {circumflex over (N)}_(nca) is the equivalent number of coherent packets; Ŵ is the equivalent search window; and {tilde over (Z)} is a mean power estimate value calculated for a first 80 256-chip power estimate outputs N, by the equation: $\hat{Z} = \frac{{\sum\limits_{n = 1}^{N}x_{n}^{2}} + y_{n}^{2}}{N}$ where: x_(n) and y_(n) are the respective in-phase and quadrature input samples, and for following samples modified according to the following infinite impulse filter (IIR) filter: {tilde over (Z)} _(k)=(1−α){tilde over (Z)} _(k-1) +α{tilde over (Z)} where α= 1/16.
 5. The method of claim 4, wherein: said constant C_({circumflex over (N)}) _(nca) _(,Ŵ) is selected from the table: W = 512 W = 1024 N_(nca) = 1 N_(nca) = 2 N_(nca) = 4 N_(nca) = 8 N_(nca) = 1 N_(nca) = 2 N_(nca) = 4 N_(nca) = 8 5.104674 7.769179 12.155090 9.756381 5.221810 7.952168 12.398603 20.086897

according to the selected values of W and N_(nca). 